Circuit arrangement for a wideband mixer with predistortion and high linearity

ABSTRACT

Circuit arrangement for a wideband mixer ( 1, 101, 201 ) with a multiplicative mixing stage ( 2 ) which exhibits a carrier frequency input ( 3, 4 ) for coupling in a differential carrier frequency signal (LOP, LON), a mixing stage input ( 5, 6 ) for coupling in a predistorted differential input signal (LFPD, LFND), and an output ( 9, 10 ) for coupling out a differential output signal (OUTP, OUTN) which is generated from the differential carrier frequency signal (LOP, LON) and the predistorted differential input signal (LFPD, LFND) by multiplicative mixing, the predistorted differential input signal (LFPD, LFND) being generated from a differential input signal (LFP, LFN) by means of a quadratic predistortion and a linear predistortion.

The present invention relates to a circuit arrangement for a widebandmixer with predistortion and high linearity. The invention also relatesto a method for wideband mixing.

Mixers are needed for frequency conversion in transmitters and receiversand, therefore, belong to the important components for wirelesstransmission systems. An ideal mixer can be implemented by means of amultiplier. This multiplies a local-oscillator signal by an input signalto be converted to form an output signal.

Conventional mixers are operated within a relatively narrowcarrier-frequency range—such as, for example, in WLAN (wireless localarea network) applications around 20 MHz. Recently, however,particularly wideband applications are needed in frequency bands from 3to 10 GHz, so-called UWB (ultra wide band) applications.

FIG. 1 shows a basic modulator or mixer topology.

Two single-sideband mixing branches are provided, the signals OUTU, OUTLof which are combined by means of an adder A1 to form the modulatedoutput signal MOUT. In the branch for the upper sideband, digital inputsignal data DLF1, which are converted into the analog signal LF1 by adigital/analog converter DA1, are generated from a ROM memory SINROMwhich contains sinusoidal data, in dependence on a control signal S1.The analog signal LF1 is filtered by a low-pass filter LPF1 to form thesignal LH1′ to be converted.

The analog signal LH1′ to be converted is mixed with a sinusoidalcarrier sequence signal with the angular frequency ω_(LO) in amultiplicative mixing stage M1 to form the output signal OUTU for theupper sideband.

Analogously, digital data DLF2, an analog conversion signal LF2 for thelower sideband and, by means of a filtered conversion signal LF2′ and acosinusoidal carrier signal for the lower sideband with the angularfrequency ω_(LO), an output signal OUTL are generated in the branch forthe lower sideband by means of a ROM memory COSROM containingcosinusoidal data, a second digital/analog converter DA2, a low-passfilter LPF2 and a multiplicative mixing stage M2 by means of a controlsignal S2.

A possible multiplicative mixing stage is represented by the so-calledGilbert cell. Such a multiplicative mixing stage according to the priorart is described, for example, in U. Tietze, CH. Schenk:Halbleiter-schaltungstechnik (Semiconductor circuit technology), Edition12, Springerverlag Berlin, Heidelberg, New York, ISBN 3-540-42849-6.

FIG. 2 shows a corresponding Gilbert cell for multiplicatively mixing adifferential input signal LFP, LFN with a differential carrier frequencysignal LOP, LON to form a differential output signal OUTP, OUTN.

Accordingly, a first resistor R1, the controlled path of a firsttransistor T1, the controlled path of a second transistor T2 and thecontrolled path of a current source transistor T3 are provided in seriesbetween a first supply voltage potential VDD and a second supply voltagepotential VSS.

A second resistor R2 and the controlled path of a fourth transistor T4are provided between the first supply voltage potential VDD and thecontrolled path of the second transistor T2. A bias potential BIAN isapplied to a gate terminal of the current source transistor T3.

The controlled path of a fifth transistor T5 and the controlled path ofa sixth transistor T6 are connected between the second resistor R2 andthe controlled path of the current source transistor T3. Furthermore,the controlled path of a seventh transistor T7 is connected between thefirst resistor R1 and the controlled path of the sixth transistor T6.

A first component LFP of the differential input signal is connected tothe gate terminal of the sixth transistor T6 and the second componentLFN of the input signal is connected to the gate terminal of the secondtransistor T2.

A first component LOP of the carrier frequency signal is connected tothe gate terminal of the first transistor T1 and the gate terminal ofthe fifth transistor T5. The second component LON of the differentialcarrier frequency signal is connected to the gate terminal of the fourthtransistor T4 and the gate terminal of the seventh transistor T7.

The first component OUTP of the differential output signal is picked upbetween the first resistor and the controlled path of the firsttransistor T1. The second component OUTN of the output signal is pickedup between the second resistor and the controlled path of the fourthtransistor T4.

The resistors R1 and R2 act as load resistors. Such a Gilbert cellaccording to FIG. 2 only has inadequate characteristics of linearity forUWB applications. This leads to harmonic distortions, in particularbecause the output characteristic of the differential pair oftransistors T1, T5, T7, T4 is quadratic.

When they are constructed as NMOS transistors, transistors T1-T7 operatein the inversion region so that the voltages dropped across the loadresistors R1, R2 are proportional to the square root of the inputvoltage which is coupled in by the differential signal LFP, LFN.

With ideal components, this nonlinearity can be largely compensated forby a quadratic predistortion of the input signal LFP, LFN. For thispurpose, NMOS diodes can be used, for example, which are connectedbetween the first supply voltage VDD and the respective gate terminalsof the transistors T2 and T6. As a result, a quadratic predistortedinput signal component in each case occurs which can compensate for thesquare root nonlinearity of the mixing stage.

In this arrangement, however, accurate matching of the two diodes isvery difficult, particularly when the mixing stage is constructed inMOS. A mismatch of such predistortion diodes easily leads to additionalcomponents of the carrier frequency in the output signal of such amixing stage.

It is, therefore, an object of the present invention to create a circuitarrangement for a wideband mixer with predistortion and high linearitywhich can be used in UWB applications and is tolerant to mismatches inthe predistortion.

It is also an object of the present invention to create a wideband mixerwhich offers a greater spurious-interference-free dynamic range comparedwith mixers with quadratic predistortion.

According to the invention, this object is achieved by a circuitarrangement for a wideband mixer according to claim 1 and by a methodfor mixing a carrier frequency signal and an input signal having themethod steps according to claim 14.

Accordingly, a circuit arrangement for a wideband mixer with amultiplicative mixing stage is provided which exhibits a carrierfrequency input for coupling in a differential carrier frequency signal,a mixing stage input for coupling in a predistorted differential inputsignal and an output for coupling out a differential output signal. Inthis arrangement, the differential output signal is generated from thedifferential carrier frequency signal and the predistorted differentialinput signal by multiplicative mixing. The predistorted differentialinput signal is generated from a differential input signal by means of aquadratic predistortion and a linear predistortion.

According to the method according to the invention for mixing a carrierfrequency signal and an input signal, the following method steps areprovided:

-   (a) linear predistorting and quadratic predistorting of the input    signal for generating a predistorted input signal; and-   (b) multiplicative mixing of the predistorted input signal with the    carrier frequency signal to form an output signal.

The basic concept of the invention consists in performing a linearpredistortion in addition to a quadratic predistortion of the inputsignal to be converted.

As a result, the influence of a mismatch of components in the signalpath, which generate quadratic predistortion, is reduced. Furthermore,output signal components which exhibit the frequency of the carriersignal are reduced. Since artifacts of the carrier frequency in theoutput signal are reduced by the linear predistortion in comparison withthe signal components to be transmitted, particularly in the upper andlower sideband and with respect to higher harmonic conversion products,a mixer according to the invention provides a particularly largespurious-interference-free dynamic range.

The linear signal component which is achieved by the additionalpredistortion according to the invention is then adjusted in such amanner that, for example, the difference of the transmitted energy bythe carrier and the transmitted energy in a relevant sideband becomesmaximum. In particular, future UWB applications specify particular masksfor the spectral densities in which the transmitted energy is plottedwith respect to frequency, of the mixer used. In this arrangement, theso-called spurious-interference-free dynamic range of the mixer isspecified which is particularly high for the circuit arrangementaccording to the invention.

Moreover, the wideband mixer according to the invention meets highlinearity requirements over a very large frequency range.

The differential input signal advantageously exhibits a first componentwhich is coupled to a first input terminal, and exhibits a secondcomponent which is coupled to a second input terminal. In thisarrangement, a first resistor is connected between the first inputterminal and a first supply voltage potential and a second resistor isconnected between the second input terminal and the first supply voltagepotential for the purpose of linear predistortion. A first component ofthe predistorted differential input signal can then be picked up betweenthe first resistor and the first input terminal, and a second componentof the predistorted differential input signal can be picked up betweenthe second resistor and the second input terminal. By connecting theresistor, a linear signal component is impressed on the respective inputsignal components.

The first and second resistor is particularly advantageously anadjustable resistor. As a result, the linear predistortion can beflexibly adapted.

According to an embodiment of the circuit arrangement according to theinvention, the first and second resistors are switchable and furtherswitchable resistors connected in parallel with these are provided.

According to a preferred embodiment, the circuit arrangement accordingto the invention exhibits a first and a second transistor having in eachcase a controlled path and a gate terminal, the controlled path of thefirst transistor being connected between the input terminal and thefirst supply voltage potential. In this arrangement, the controlled pathof the second transistor is connected between the second input terminaland the first supply voltage potential and the gate terminals areconnected to one another. The gate terminals are connected to adistortion potential.

Changing the distortion potential changes the characteristics of thecontrolled paths in such a manner that the extent of the quadraticpredistortion which occurs due to the controlled paths is alsoadjustable.

In an advantageous embodiment, the distortion potential is the firstsupply potential. The first and second transistors are then connected asdiodes. These diodes provide exactly the desired quadraticpredistortion.

In an alternative embodiment, a first diode which is connected betweenthe first input terminal and the first supply voltage potential isprovided and a second diode, which is connected between the second inputterminal and the first supply potential is provided.

In a preferred embodiment, the multiplicative mixing stage exhibits aGilbert cell.

In a preferred development of the circuit arrangement, a predistortioncontrol is also provided, having a test signal output for outputting adifferential test signal to the input terminals of the circuitarrangement, with a test signal input for coupling in the differentialoutput signal of the mixing stage, and with at least two control outputsfor outputting adjustment signals for adjusting the first and secondresistor.

The linear predistortion can be optimally adjusted by adjusting theresistance values of the resistors by coupling in test signals at theinput of the circuit arrangement and, at the same time, evaluating theoutput signal generated.

The circuit arrangement is advantageously constructed in MOS technology.Current-saving components which are simple to produce are indicatedparticularly in the case of UWB applications. This can be achieved in aparticularly simple manner in MOS technology.

The resistors for linear predistortion are preferably dimensioned insuch a manner that the spurious-interference-free dynamic range of theoutput signal of the mixer according to the invention is maximum with adifferential input signal with predetermined frequency.

The resistors can be adjusted or dimensioned, for example, by means ofthe predistortion control or determined by simulations even beforeproduction.

In a particularly preferred embodiment, the resistors for linearpredistortion are dimensioned in such a manner that over a predeterminedrange of carrier frequencies the signal energy of an output signal isequal to the signal energy of a third harmonic of the input signal inthe output signal at the frequency of the carrier signal. Since thelinear predistortion, in particular, generates third harmonics as abyproduct, a particularly large spurious-interference-free dynamic rangecan be achieved by means of this adjustment of the predistortion.

Further advantageous embodiments and developments of the invention aresubject-matter of the subclaims and of the subsequent description andreferring to the figures.

In the text which follows, the invention will be explained in greaterdetail with reference to the diagrammatic figures and exemplaryembodiments. In the figures:

FIG. 1 shows a single-sideband mixer architecture according to the priorart;

FIG. 2 shows a Gilbert cell according to the prior art;

FIG. 3 shows a first embodiment of the circuit arrangement according tothe invention for a wideband mixer;

FIG. 4 shows a spectral representation of the conversion products of amixer according to the invention and of a conventional mixer;

FIG. 5 shows a second embodiment of the circuit arrangement according tothe invention;

FIG. 6 shows a third embodiment of the circuit arrangement according tothe invention; and

FIG. 7 shows a preferred development of the circuit arrangementaccording to the invention.

FIGS. 1 and 2 have already been described in the introduction.

FIG. 3 shows a first embodiment of the circuit arrangement according tothe invention for a wideband mixer 1.

A multiplicative mixing stage 2 is provided which is constructed, forexample, as a Gilbert cell. The multiplicative mixing stage 2 exhibits acarrier frequency input 3, 4 into which the differential carrierfrequency signal LOP, LON is coupled.

The differential carrier signal exhibits a first component LOP which isconnected to the first carrier signal terminal 3, and the differentialcarrier signal exhibits a second component LON which is connected to asecond carrier signal terminal 4.

The multiplicative mixing stage 2 exhibits a mixing stage input 5, 6 forcoupling in a predistorted differential input signal LFPD, LFND. In thisarrangement, a first component LFPD of the predistorted differentialinput signal is connected to a first mixing stage input terminal 5, anda second component LFD of the differential predistorted input signal isconnected to a second mixing stage input terminal 6.

The differential input signal LFP, LFN is coupled in at an input 7, 8 ofthe mixer 1 according to the invention. In this arrangement, a firstcomponent LFP of the differential input signal is connected to a firstinput terminal 7 and a second component LFN of the differential inputsignal is connected to a second input terminal 8.

The differential output signal OUTP, OUTN is output at an output 9, 10.In this arrangement, a first component OUTP of the differential outputsignal can be picked up at an output terminal 9 and a second componentOUTN of the differential output signal can be picked up at a secondoutput terminal 10.

The multiplicative mixing stage 2 exhibits a first resistor 11, a secondresistor 12, a current source transistor 13 with a controlled path and agate terminal 14, a first, second, third, fourth, fifth and sixthtransistor 15, 17, 19, 21, 23, 25 having in each case a controlled pathand a gate terminal 16, 18, 20, 22, 24, 26.

The first resistor 11, the controlled path of the first transistor 15,the controlled path of the sixth transistor 25 and the controlled pathof the current source transistor 13 are connected serially between afirst supply voltage potential VDD and a second supply voltage potentialVSS.

The second resistor 12, the controlled path of the second transistor 17and the controlled path of the fifth transistor 23 are connectedserially between the first supply voltage potential VDD and thecontrolled path of the current source transistor 13.

A first node 27 is provided between the first resistor 11 and thecontrolled path of the first transistor 15. A second node 28 is providedbetween the second resistor 12 and the controlled path of the fourthtransistor 21.

The controlled path of the second transistor 17 is connected between thesecond node 28 and the controlled path of the fifth transistor 23. Thecontrolled path of the third transistor 19 is connected between thefirst node 27 and the controlled path of the fifth transistor 23.

The gate terminal 16 of the first transistor 15 and the gate terminal 18of the second transistor 17 are connected to the first carrier frequencyterminal 3. The gate terminal 20 of the third transistor and the gateterminal 22 of the four transistor is connected to the second carrierfrequency terminal 4.

The first component OUTP of the output signal can be picked up at thefirst node 27 and is connected to the first output terminal 9. Thesecond component OUTN of the output signal can be picked up at thesecond node 28 and is connected to the second output terminal 10.

The gate terminal 24 of the fifth transistor 23 is coupled to the firstmixing stage input terminal 5, and the gate terminal 26 of the sixthtransistor 25 is connected to the second mixing stage input terminal 6.

In each case, an output signal current IOUTP, IOUTN, flows via the firstand second resistor 11, 12, as a result of which the respective signalvoltage of the output signal OUTP, OUTN is generated.

Between the first input terminal 7 and the first supply voltagepotential VDD, a third resistor 29 and the controlled path of a seventhtransistor 30 are connected in parallel.

Between the first supply voltage potential VDD and the second inputterminal 8 of the mixer 1, a fourth resistor 31 and the controlled pathof an eighth transistor 32 are connected in parallel. The gate terminals33, 34 of the seventh and eighth transistors 30, 32 are applied to adistortion potential BPRED. The gate terminal 14 of the current sourcetransistor 13 is applied to a BIAS potential BIAN.

As has already been mentioned in the introduction, the multiplicativemixing stage 2 (Gilbert cell) does not operate in a linear manner, i.e.the output currents IOUTP, IOUTN are proportional to the square root ofthe input voltage of the differential input signal LFP, LFN.

The input signal LFP, LFN is quadratically predistorted by means of theseventh and eighth transistor 30, 32, so that at least a part of thenonlinearity due to the mixing stage is compensated for. However, sinceit is difficult in production to make the transistors 30, 32 completelyidentical, a mismatch can easily occur. A mismatch of the seventh andeighth transistors 30, 32 causes interference mainly of the carrierfrequency signal LOP, LON with the output signal OUTP, OUTN.

The additional linear predistortion which is also generated by the thirdand fourth resistor 29, 31, effectively reduces the interference of thecarrier frequency signal LOP, LON with the output signal OUTP, OUTN.

The amount of linear predistortion is set by choosing the resistancevalues of the third and fourth resistor 29, 31 and the amount ofquadratic predistortion is set by changing the distortion potentialBPRED which is applied to the gate terminals 33, 34 of the seventh andeighth transistor 30, 32.

FIG. 4 shows a spectral representation of the conversion products of amixer according to the invention as shown, for example, in FIG. 3.

FIG. 4 shows the output spectrum of a mixer only using quadraticpredistortion by means of MOS diodes (PDD spectral lines) and thespectrum of a mixer according to the invention using quadratic andlinear predistortion by means of diodes and resistors (PRD spectrallines).

The frequency of the carrier frequency signal has here been chosen as6.18 gigahertz and the input signal as 660 megahertz.

The spurious-interference-free dynamic range of the mixer using onlyquadratic predistortion is obtained from the difference between theenergies (here represented as degree of amplification in decibels asratio of the input voltage to the output voltage), resulting in aspurious-interference-free dynamic range of ΔPDD=28.25 decibels. This isobtained from the difference between the degree of amplification of theupper sideband USB and the degree of amplification of the carrier signalC.

The dynamic range ΔPRD of the mixer with linear and quadraticpredistortion is about 33.9 decibels and is thus greatly improved.

The linear predistortion can lead to an amplification of the thirdharmonic (or third-order harmonic, respectively) of the input signal inthe output spectrum of the mixer.

In the example chosen here, the linear predistortion or the resistancevalue, respectively, is set in such a manner that the degree ofamplification of the third harmonic 3HM is equal to that of the carriersignal C. In this manner, a compromise between the generation of thethird harmonic 3HM and an extended dynamic range ΔPRD can be achieved bysetting the resistances or linear distortion, respectively. The spectrumaccording to FIG. 4 also shows the lower sideband LSB.

FIG. 5 shows a second embodiment of the circuit arrangement according tothe invention for a wideband mixer.

The second embodiment essentially exhibits the same components as theembodiment according to FIG. 3 but the predistortion potential isidentical with the first supply voltage potential VDD.

The seventh and the eighth transistor 30, 32 is thus in each caseconnected as a diode. Furthermore, a ninth and tenth transistor 35, 37having in each case a controlled path and a gate terminal 36, 38 and asecond current source transistor 39 having a controlled path and a gateterminal 40 are provided.

In the second embodiment of the mixer 101, the first component LFP ofthe differential input signal is coupled to the gate terminal 36 of theninth transistor 35. The second component LFN of the differential inputsignal is coupled to the gate terminal 38 of the tenth transistor 37.

The controlled path of the ninth transistor 35 and the controlled pathof the second current source transistor 39 is connected between thethird resistor 29 and, respectively, the controlled path of the seventhtransistor 30 and the second supply voltage potential VSS.

The controlled path of the tenth transistor 37 is connected between thefourth resistor 31 and, respectively, the controlled path of the eighthtransistor 32 and the controlled path of the second current sourcetransistor 37.

FIG. 6 shows a third embodiment 201 of the mixer according to theinvention.

A multiplicative mixing stage 2 is provided which exhibits an input 3, 4for the carrier frequency signal LON, LOP, an output 9, 10 for thedifferential output signal OUTP, OUTN, and a first mixing stage inputterminal 5 for the first component LFPD of the predistorted input signaland a second mixing stage input terminal 6 for the second component LFNDof the predistorted input signal.

The first component LFP of the differential input signal is coupled to afirst input terminal 7 and the second component LFN of the differentialinput signal is coupled to a second input terminal 8.

A first transistor 41 with a controlled path and a gate terminal 43 anda second transistor 42 with a controlled path and a gate terminal 44 areprovided. The controlled path of the first transistor 41 is connectedbetween a first supply voltage potential VDD and the first mixing stageinput terminal 5. The gate terminal 43 of the first transistor 41 isconnected to the first supply voltage potential VDD.

The controlled path of the second transistor 42 is connected between thefirst supply voltage potential VDD and the second mixing stage inputterminal 6. The gate terminal 44 of the second transistor 42 isconnected to the first supply voltage potential VDD.

Three resistors 48, 49, 50, which can be connected via switches 45, 46,47 are arranged in parallel so that they can be connected to thecontrolled path of the first transistor 41.

Three further resistors 54, 55, 56, which can be connected via switches51, 52, 53, are connected in parallel so that they can be connected tothe control path of the second transistor 42.

The two transistors 41, 42 are again connected as diodes and generatethe quadratic predistortion of the differential input signal LFP, LFN.

Various linear predistortions can be set by combining the connectableresistors 48, 49, 50, 54, 55, 56 in various ways. As a result, theoptimum linear predistortion can be easily adjusted in order to achievethe greatest possible spurious-interference-free dynamic range.

FIG. 7 shows a preferred development of the mixer 202 according to theinvention.

A multiplicative mixing stage 2 is again provided, having carrier signalterminals 3, 9, mixing stage input terminals 5, 6 and output terminals9, 10. To provide quadratic predistortion, two diode-connectedtransistors 41, 42 are provided between the mixing stage input terminals5, 6 and the first supply voltage potential VDD as in FIG. 6.

The controlled path of the first transistor 41 is connected in parallelwith an adjustable or programmable resistor 57. The controlled path ofthe second transistor 42 is connected in parallel with a secondadjustable or programmable resistor 58.

Furthermore, a predistortion control 60 is provided which exhibits atest signal output 61, 62 for outputting a differential test signalTLFP, TLFN to the input terminals 7, 8. The predistortion control 60 hasa test signal input 63, 64 to which the differential output signal OUTP,OUTN of the mixing stage 2 is coupled. The predistortion control 60 alsohas at least 2 control outputs 65, 66 for outputting the adjustmentsignals CTR1, CTR2 for adjusting the first and second resistor 57, 58.

The predistortion control 60 provides, for example, a differential testsignal TLFP, TLFN with constant frequency which is coupled to theterminals 7, 8.

At the same time, the predistortion control 60 receives the resultantoutput signal OUTP, OUTN of the multiplicative mixing stage 2.

The predistortion control 60 controls the resistance values of the firstand second resistor 57, 58 via the adjustment signals CTR1, CTR2, insuch a manner that the spurious-interference-free dynamic range becomesmaximum. This can be done, for example, as explained in FIG. 4 by meansof the spectral data representation. The predistortion control 60 canboth be integrated in the mixer 202 according to the invention or act asan external device. In this case, the programmable resistors 57, 58 canbe programmed accordingly, for example during production.

The present circuit arrangement for a wideband mixer, therefore,provides excellent linearity over a wide frequency range, is preferablyconstructed in CMOS technology as a result of which the currentconsumption is low, supplies an improved spurious-interference-freedynamic range compared with the prior art and can thus be used in futureultra wideband applications.

LIST OF REFERENCE DESIGNATIONS

-   S1, S2 Control signal-   DLF1, DLF2 Input signal data-   LF1, LF2, LF1′, LF2′ Analog signal-   OUTU, OUT1, MOUNT Output signal-   SINROM, COSROM ROM memory-   DR1, DR2 Digital/analog converter-   LPF1, LPF2 Low-pass filter-   M1, M2 Multiplicative mixing stage-   A1 Adder-   LOP, LON Carrier frequency signal-   OUTP, OUTN Output signal-   T1-T6 Transistor-   T3 Current source transistor-   R1, R2 Resistor-   BIAN Bias potential-   VDD, VSS Supply voltage potential-   IOUTP, IOUTN Output signal current-   BPRED Distortion potential-   LSB Lower sideband-   USB Upper sideband-   PDD, PRD Spectral lines-   C Carrier signal-   ΔPDD, ΔPRD Spurious-interference-free dynamic range-   3HM Third harmonic-   LPFPD, LFND Predistorted input signal-   CTR1, CTR2 Adjustment signal-   TLFP, TLFN Test signal-   1 Mixer-   2 Multiplicative mixing stage-   3, 4 Carrier signal terminal-   5, 6 Mixing stage input terminal-   7, 8 Input-   9, 10 Output-   11, 12 Resistor-   13 Current source transistor-   14 Gate terminal-   17, 19, 21, 23, 25 Transistor-   18, 20, 22, 24, 26 Gate terminal-   31 Resistor-   32 Transistor-   34 Gate terminal-   37 Transistor-   38 Gate terminal-   39 Current source transistor-   40 Gate terminal-   41 Switching transistor-   43, 44 Gate terminal-   45, 46, 47, 51, 52, 53 Switch-   48, 49, 50, 54, 55, 56 Resistor-   57, 58 Programmable resistor-   60 Predistortion control-   61, 62 Test signal output-   63, 64 Test signal input-   65, 66 Control output-   101 Mixer-   202 Mixer

1-15. (canceled)
 16. A circuit arrangement for a wideband mixer,comprising: a multiplicative mixing stage having a carrier frequencyinput, a differential signal input, and a differential mixing stateoutput; a predistortion circuit coupled to the differential signalinput, and further configured to receive at least a part of adifferential input signal, the predistortion circuit configured togenerate a differential output signal using quadratic predistortion andlinear predistortion, the predistortion circuit operable to provide thedifferential output signal to the differential signal input of themultiplicative mixing stage.
 17. The circuit arrangement as claimed inclaim 16, wherein the predistortion circuit further includes a firstpredistortion section having a first input terminal configured toreceive a first component of the differential input signal and a secondpredistortion section having a second input terminal configured toreceive a second component of the differential signal.
 18. The circuitarrangement as claimed in claim 17, wherein the first predistortionsection includes a first resistor coupled between a first supply voltagepotential and the first input terminal, and the second predistortionsection includes a second resistor coupled between the second inputterminal and the first supply voltage potential.
 19. The circuitarrangement as claimed in claim 18, wherein at least the first resistorcomprises an adjustable resistor.
 20. The circuit arrangement as claimedin claim 18, wherein the first predistortion section further comprises aplurality of resistors including the first resistor, each of theplurality of resistors being switchably connected in parallel betweenthe first supply voltage terminal and the first input terminal.
 21. Thecircuit arrangement as claimed in claim 17, wherein the predistortioncircuit includes a first transistor having a controlled path connectedbetween the first input terminal and a first supply voltage potential,and a second transistor having a controlled path connected between thesecond input terminal and the first supply voltage potential, and thefirst and second transistors having gate terminals connected to oneanother and connected to a distortion potential.
 22. The circuitarrangement as claimed in claim 21, wherein the distortion potential isthe first supply voltage potential.
 23. The circuit arrangement asclaimed in claim 16, wherein the predistortion circuit further includesat least one diode configured to generate the quadratic predistortion.24. The circuit arrangement as claimed in claim 16, wherein the whereinthe multiplicative mixing stage comprises a Gilbert cell.
 25. Thecircuit arrangement as claimed in claim 19, further comprising apredistortion control circuit, the predistortion control circuitincluding: a test signal output configured to provide a test signal tothe first input terminal and the second input terminal; a test signalinput coupled to the differential mixing state output; and at least onecontrol output configured to provide adjustment control signals to theadjustable first resistor.
 26. The circuit arrangement as claimed inclaim 16, wherein the circuit arrangement is constructed in MOStechnology.
 27. The circuit arrangement as claimed in claim 16, whereinthe linear predistortion is selected in such a manner that thespurious-interference-free dynamic range of an output signal of themultiplicative mixing stage is maximum for a differential input signalhaving a predetermined frequency.
 28. The circuit arrangement as claimedin claim 16, wherein the multiplicative mixing stage comprises asingle-sideband mixing stage.
 29. The circuit arrangement as claimed inclaim 16, wherein the linear predistortion is selected such that, over apredetermined range of carrier frequencies, the signal energy of anoutput signal is equal to the signal energy of a third harmonic of thedifferential input signal within the output signal at the frequency ofthe carrier frequency signal.
 30. A method for mixing a carrierfrequency signal and an input signal, comprising: (a) effecting linearpredistortion and quadratic predistortion of the input signal in orderto generate a predistorted input signal; and (b) mixing the predistortedinput signal with the carrier frequency signal to form an output signal.31. The method as claimed in claim 30, wherein the linear component ofthe predistortion is selected in such a manner that thespurious-interference-free dynamic range of the output signal is maximumwith an input signal which exhibits a predetermined frequency.
 32. Themethod as claimed in claim 30, wherein step a) further compriseseffecting linear distortion at least in part using a resistive circuithaving at least one resistor and a circuit resistance.
 33. The method asclaimed in claim 32, wherein step a) further comprises effecting lineardistortion using a resistive circuit having an adjustable circuitresistance.
 34. A circuit arrangement for a wideband mixer, comprising:a mixing stage having a carrier frequency input, a differential signalinput, and a differential mixing state output; a predistortion circuitcoupled to the differential signal input, and further configured toreceive at least a part of a differential input signal, thepredistortion circuit configured to generate a differential outputsignal using quadratic predistortion and linear predistortion, thepredistortion circuit including a resistive circuit having a resistancevalue to generate the linear predistortion, the predistortion circuitoperable to provide the differential output signal to the differentialsignal input of the mixing stage.
 35. The circuit arrangement accordingto claim 34, wherein the predistortion circuit further comprises anonlinear circuit configured to generate the quadratic predistortion,the nonlinear circuit comprising at least one of the group consisting ofa transistor and a diode.